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Results: The algorithm described above was applied to test
sequences containing a significant amount of undesirable global
motion. Interlaced-to-progressiveconversion was implemented by
dropping every other field and vertically upsampling the retained
fields by 2:l to restore ITUR Rec. 601 spatial resolution. No antialiasing pre-filtering was applied in this case. Blocks of 64 x 64
pixels were used yielding a field of 11 x 9 = 99 motion vectors and
a total of 10 x 9 + 11 x 8 = 178 block pairs to be processed. The
accuracy of estimation was 11100 of a degree for the angle of rotation, 1i1000 for the magnitude of scale changes and 114 pixel for
Undesirable global motion in the test sequences under consideration is due to vibrations resulting from the use of a portable camera. Fig. 3 shows the luminance component of the first frame of
test sequence ‘rescue’. In the middle of the Figure, the difference
between the first and the second frame shows evidence of significant amounts of global motion. At the bottom of the Figure, the
result of motion compensating the second frame with respect to
the first, according to the estimated global motion, is shown. The
estimated parameters for this pair of frames were 4 = - 0.86
degrees, a = 1.003, zx = 8.000 pixels and z, = -5.5 pixels. Using
the PSNR as a measure of frame difference, this was measured at
19.25dB for the original pair and at 30.85dB after motion compensation. To prevent edge effects affecting the PSNR measurements, only the centred 640 x 500 pixels were used in this
computation. The residual frame difference after motion compensation is due to scene motion as well as global motion containing
components other than pure rotation, change of scale and translation.
Conclusion: A simple algorithm for the estimation of global
motion parameters from sparse translational vector fields was
described. An important feature of the proposed technique is its
low complexity which makes it suitable for real-time implementations. Another key advantage is compatibility with motion estimation tools that already exist within internationally standardised
video compression algorithms employing motion-compensated
prediction. Our results show that accurate estimates of global
motion parameters can be obtained from sequences which contain
complex scene motion as well as global motion due to hand-held
camera vibrations. In this case, global motion compensation using
the computed estimates offers significant benefits in both objective
and subjective quality owing to the drastic reduction of camera
Finally, the proposed technique will provide reliable estimates
only when the majority of raw motion measurements correspond
to areas of the picture whose depth map is relatively smooth (such
as non-complex backgrounds). This condition is approximately
met by many typical scenes.
Fully differential current operational
Sibum Jun and Dae Mann Kim
A fully differential, wide bandwidth and good CMRR current
operational amplifier is presented. The proposed circuit employs a
tapped cascode current mirror as an input stage and simple
common source stages as an outpnt stage. To enhance the
common-mode rejection ratio (CMRR), a hgh impedence node is
implemented by two cross-coupled simple current mirrors.
Introduction: Current-mode operational amplifier (COA) has
advantages over its voltage-mode counterpart, such as higher
operating frequency and dynamic range, etc. Various designs were
reported in literature [l - 51; among them, the current steering
approach [1] is novel and shows a superior performance, but was
based on single-ended input operation. The advantages of the fully
differential designs over single-ended versions are obvious. In this
Letter, we propose a very simple, fully differential design of a
COA having good common-mode rejection ratio (CMRR), wide
bandwidth and low power consumption.
Fig. 1 Proposed circuit
Proposed circuit: The proposed circuit shown in Fig. 1 is organised
and Mp21Mp4),
as two tapped cascode current mirror (Mp,lMp3
and two simcross-coupled simple mirrors (Mn,/Mdand Mn3iMn4),
ple common-source stages (Mnsand MnJ. The input signals, Ap
and An,are injected into the tapped internal nodes of the cascode
current mirrors. Since the tapped cascode current mirrors are basically common-gate amplifiers with constant bias current I,,,,,the
injected signals are delivered, nearly intact, to the following crosscoupled simple mirrors. The input resistance r,,, and output resistance r,,, of the input stage are:
0 IEE 1998
8 October 1997
Electronics Letters Online No: 19980064
T. Vlachos (BBC Research and Development, Kingswood Warren,
Tadworth, Surrey KT20 6NP, United Kingdom)
The author is currently with Centre for Vision, Speech and Signal
Processing, University of Surrey, Guildford, Surrey GU2 5XH, United
WATANABE, H.: ‘Two-stage motion compensation using adaptive
global MC and local affine MC’, IEEE Trans. Circuits Syst. Video
Technol., 1997, 7, (l), pp. 75-85
VLACHOS, T., and THOMAS, G.: ‘Motion estimation for the correction
of twin-lens telecine flicker’. Proc. IEEE-ICIP, 1996, Vol. 1, pp.
109-1 12
and ISHII, H.: ‘Electronic image
stabilisation system for video cameras and VCRS’, S M P T E J.,
1992, pp. 6 6 7 5
HOETTER, M.: ‘Differential estimation of the global motion
parameters zoom and pan’, Signal Process., 1989, 16, pp. 249-265
wu, s.F., and KITTLER, J.: ‘A differential method for simultaneous
estimation of rotation, change of scale and translation’, Signal
Process.: Image Commun., 1990, 2, pp. 69-80
Rec. H.262, ‘Generic coding of moving pictures and associated
audio: Video’. MPEG-2 Video Compression Specification, ISO/
IEC 13818-2, 1994
THOMAS, G.A.: ‘Television motion measurement for DATV and
other applications’. BBC Res. Dept. Rep., No. 1987/11
rout 2[rop r o p ( l gmprop)]
where gmpand rOpare the small-signal transconductance and drain
resistance of PMOS transistors M,, to Mp4,respectively.
The cross-coupled smple mirrors are adopted for increasing the
CMRR. This circuit provides a high incremental resistance (rh)
for the differential-mode signals, while having a small resistance
(rc,) for the common-mode ones. Assuming matched identical
devices, r,, and r, can be shown to be equal to
rdm 21 T o n
where g,, and v,, are the small-signal transconductance and drain
resistance of NMOS transistors M,, to M,,, respectively. Hence,
the injected differential input current leads to a large incremental
voltage swing, while the common-mode input current has a much
smaller amplitude compared to the differential one. It should be
mentioned here that the proposed COA operates very similarly to
the steered COA [I], at least in a differential manner, where the
current mirrors are excluded intentionally to achieve a higher
speed. Since the cross-coupled simple mirrors cancel each other’s
transconductance values, no current mirroring operation exists in
the differential-mode signal path. Current-mirroring is valid only
on the common-mode signal path. If the normal current source
8th January 1998
Vol. 34
No. I
which is not cross-coupled is employed, r, has a similar order of
magnitude to rd,. This leads to a poor CMRR, unless an additional common-mode feedback loop is introduced.
ydm can be further increased by slightly oversizing M,, and M,,,
compared to M,, and Mn4.However, it depends on process tolerance, which is difficult to predict and is therefore excluded in this
Finally, the voltages on the high impedence nodes are converted
again to the current by the small-signal transconductance (g") of
Mn5and Mn6.The overall differential current gain AIdmand the
common-mode current gain A,, of the proposed circuit are:
pensation capacitors. The CMRR obtained is also shown in Fig.
2. As mentioned, the CMRR has a magnitude nearly equal to the
differential-modeopen loop gain. A further enhanced CMRR will
be obtained if an additional common-mode feedback control is
introduced. The proposed COA was simulated in a closed loop
unity gain buffer. Fig. 4 shows the transient response to a step
input of flOpA. The 1% settling time is found to be 7.lns. The
input and output resistances of the COA are 21 and 597kQ
respectively. Finally the static power dissipation was < 860pW.
Since the common-mode gain is about one if gmm-- gmn,
CMRR of the proposed COA is nearly equal to A,,,.
The dominant pole, f , is determined by the high impedence
node and equal to:
where C, is the half of the total capacitance connected to each
high impedence node, including compensation capacitance if it
exists. Hence, the gain-bandwidth product (GBW) of the proposed
COA becomes g',,/(2:nCP).
Simulation: A SPICE simulation has been performed using the
level 28 model parameters provided for LG Semicon 0 . 6 n-well
CMOS technology with V,, = 0.771, vihp= 4.868, k, = 1 3 1 . 1 w
Vz and k, = 41.8pNV2. The supply voltage V, is 3V and the bias
is 40pA. All the devices used have the same length of
current IbLas
1p.The widths are: l o p for M,, to Mn4;2 0 p for Mn5to Mn8;
30pm for M,, to Mp8.PMOS transistors Mpsand Mp6are added to
bias Mns and Mn6,respectively. NMOS transistors Mns and Mn6
are auxiliary output devices. All body terminals of PMOS and
NMOS transistors are connected to V, and ground, respectively.
Fig. 4 Simulated transient response
Conclusions: A fully differential, wide bandwidth and good
CMRR current operational amplifier has been presented. The proposed circuit employs a tapped cascode current mirror as an input
stage and two cross-coupled simple current mirrors to enhance
CMRR. A SPICE simulation shows a 53dB differential-mode
gain, a 52dB CMRR, a 300MHz unity gain bandwidth, and a 50"
phase margin without compensation.
3 November 1997
0 IEE 1998
Electronics Letters Online No: 19980106
Sibum Jun and Dae Mann Kim (Department of Electrical Engineering,
Pohang University of Science and Technology, San 31 Hyoja-dong,
Nam-gu, Pohang, 790-784, Republic of Korea)
.. .. .. .. .. .. .. .. .. .. .. .. .. ..
.. .. .. .. .. .. .. .. .. .. ..
Fig. 2 Simulated dijjfej*ential-modeopen loop gain and C M R R
and EL-MASRY, E.I.: 'A 200MHz steered current
operational amplifier in 1 . 2 CMOS
technology', ZEEE J. SolidState Circuits, 1997, 32, (2), pp. 245-249
ABOU-ALLAM, E , and EL-MASRY, E.I.: 'High CMRR CMOS current
operational amplifier', Electron. Lett., 1994, 30, pp. 1042-1043
ZELE, R.H., LEE, s., and ALLSTOT, D.J.: 'A high gain current-mode
operational amplifier'. Proc. IEEE ISCAS, 1992, pp. 2852-2855
KAULBERG, T.: 'A CMOS current-mode operational amplifier',
IEEE J. Solid-state Circuits, 1993, 28, pp. 849-852
TOUMAZOU, c., LIDGEY, F.J., and JAIGH, D.J.: 'Analogue IC design:
The current-mode approach' (Peregrinus, London, 1990)
Design of turbo-code interleaver using
Hungarian method
A.K. Khandani
...,..., .
A method is presented for optimising the structure of the turbocode interleaver using the Hungarian method ginear sum
assignment problem). N u m e n d results are presented which show
a substantial improvement with respect to a random interleaver.
.*. ...
.. ..
. .
Fig. 3 Simulated phase characterisics
The frequency characteristics of the proposed COA are shown
in Figs. 2 and 3. It is seen that the proposed COA exhibits an
open loop gain of 53dB. The 3dB bandwidth of the COA is seen
to be 1MHz. Also, the unity gain bandwidth is found to be
300MHz, at which the phase margin is 50" without inserting com-
8th January 7998
Vol. 34
Introduction: Consider a turbo code in which the underlying recursive convolutional code (RCC) is generated by the transfer function G(d) = N(d)/D(d).We know that the impulse response of G(d)
is periodic with period p 5 2 - 1, where r is the constraint length
of the code [l]. If p = 2 - 1, the resulting impulse response is
called a maximum length sequence (MLS). This is the case for the
transfer functions used in turbo codes.
If we look at the impulse response of G(d) as a periodic
sequence, we obtain K = 2 - 1 non-zero sequences which are time
shifts of each other. We refer to these sequences as different
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